Antenna Types and Their Features

Half-Wave Dipole

The half-wave dipole antenna is the basis of many other antennas and is also used as a reference antenna for the measurement of antenna gain and radiated power density.

At the frequency of resonance, i.e. at the frequency at which the length of the dipole equals a half-wavelength, we have a minimum voltage and a maximum current at the terminations in the center of the antenna; the impedance is minimal. Therefore, we can compare the half-wave dipole antenna with a series RLC resonant circuit as given in Figure 2. For a lossless half-wave dipole antenna, the series resistance of the equivalent resonant circuit equals the radiation resistance, generally between 60 Ω and 73 Ω, depending on the ratio of its length to the diameter.

The bandwidth of the resonant circuit (or the antenna) is determined by the L-to-C ratio. A wire with a larger diameter means a larger capacitance and a smaller inductance, which gives a larger bandwidth for a given series resistance. That’s why antennas made for measurement purposes have a particularly large wire diameter.

As opposed to the (only hypothetical) isotropic radiator, real antennas such as the half-wave dipole have a more or less distinct directional radiation characteristic.

The radiation pattern of an antenna is the normalized polar plot of the radiated power density, measured at a constant distance from the antenna in a horizontal or vertical plane.

Figure shows the radiation pattern of a half-wave dipole antenna.

As the dipole is symmetrical around its axis, the three-dimensional radiation pattern rotates around the wire axis.

The isotropic gain of a half-wave dipole antenna is 2.15 dB, i.e. in the direction perpendicular to the wire axis the radiated power density is 2.15 dB larger than that of the isotropic radiator.

There is no radiation in the wire axis. The half-wave dipole produces linear polarization with the electrical field vector in line with, or in other words parallel to, the wire axis. Because the half-wave dipole is often used as a reference antenna for measurements, sometimes the gain of an antenna is referenced to the radiated power density of a half-wave dipole instead of an isotropic radiator. Also the effective radiated power ERP, which is the power delivered to an ideal dipole that gives the same radiation density as the device under test, is used instead of the EIRP. The relations Gdipole = Gisotropic − 2.15 dB and ERP = EIRP − 2.15 dB can be applied.

The half-wave dipole needs a differential feed because both of its terminations have the same impedance to ground. This can be convenient if the transmitter output or the receiver input have differential ports. A balun will be used along with the half-wave dipole in case of single ended transmitters or receivers or if an antenna switch is used.

For external ready-manufactured half-wave dipoles, the balun is visually built-in to the antenna and provides a single-ended interface.

The half-wave dipole is an electrical antenna. This means that it is easily detuned by materials with a dielectric constant larger than one within its reactive near field. If for instance the housing of a device is in the reactive near field, the housing has to be present when the antenna is matched. The human body has a large dielectric constant of approximately 75; if an electrical antenna is worn on the body or held in the hand, it can be strongly detuned.

If the antenna is built as two traces on a PCB, the dielectric constant of the PCB material has to be considered. The electrical field in the reactive near field region spreads out partially into the PCB material, partially into the surrounding air. This gives an effective dielectric constant eeff between that of the air and the PCB material /3/:

Where h is the thickness of the PCB material, w is the trace width of the dipole arms. The required length of the half-wave dipole is then:

Underneath the dipole and within the reactive near field, no ground plane is allowed.

Quarter-Wave Monopole

In many cases, the half-wave dipole is just too large. Also, the needed differential feed is often a disadvantage. If we replace one branch of the dipole antenna by an infinitely large ground plane, due to the effect of mirroring, the radiation pattern above the ground plane remains unaffected. This new structure is called a monopole antenna.

Because all the radiated power is now concentrated in the half-space above the ground plane, the gain of the monopole is 3 dB larger than the gain of the dipole.

Often a large ground plane is not feasible. The Marconi antenna replaces the (not realizable) infinitely large ground plane by several open-ended λ/4-Stubs, called the counterpoise.

A further reduction to only one stub gives a structure that looks like a bent dipole antenna. When designing a monopole antenna, the radiator should go as long as possible perpendicular to the ground stub or the ground plane. Bends close to the feeding point reduce the radiation resistance and the efficiency of the antenna.

The ideal quarter-wave monopole has a linear polarization with the vector of the electrical field in the wire axis. If the ground plane becomes unsymmetrical, the direction of polarization will be tilted towards the larger part of the ground plane, but still remains linear.

The radiation resistance of an ideal quarter-wave monopole is half of that of a dipole; depending on the ratio of length to diameter of the radiator between 30 Ω and 36.5 Ω.

Like the half-wave dipole, the quarter-wave monopole is an electrical antenna; it is influenced by the dielectric constant of the material in the reactive near field. The same formulas for the effective dielectric constant and the required length as for the half-wave dipole hold for the quarter-wave monopole.

Table 1 gives the length of half-wave dipoles and quarter-wave monopoles in free space and on a PCB for commonly used short range frequencies. For the PCB antennas, a PCB thickness of h = 1,5 mm and a trace width of w = 1 mm has been assumed; the PCB material is FR4 with εr = 4.2. This gives an effective dielectric constant of εr = 2.97.

It has to be mentioned that parasitic components such as capacitance to ground, inductance introduced by bends in the antenna as well as the influence of the package alter the antenna impedance. For monopole antennas, the ground plane is sometimes smaller than a quarter-wave length or not perpendicular to the radiator. In practice, the exact length of the dipole or the monopole has therefore to be determined by measuring the feed impedance with a vector network analyzer.

Sometimes the available space limits the length of an antenna; the antenna is made as long as the geometry permits, which can be smaller than one quarter wavelength. A monopole shorter than a quarter-wave length can be considered as a quarter-wave monopole, which is used at a frequency lower than the frequency of resonance. According to the equivalent schematic given in Figure 2, the input impedance at the frequency of operation will then be a series connection of a resistor and a capacitor. The series capacitance can be resonated out by a series inductor. A monopole antenna shorter than λ/4 with a series inductor is also referred to as a loaded stub antenna.

The radiation resistance of a loaded stub decreases with decreasing length. The smaller radiation resistance and the larger L-to-C ratio increase the quality factor and make the bandwidth smaller than for a quarter-wave monopole. Approximations for the radiation resistance of a monopole antenna are:

At the frequency of operation (i.e., resonance), the impedance of the short stub will be that of a small resistor (radiation resistance plus loss resistance) with a series capacitor. From the Smith Chart in Figure 9 we can see that matching to a 50-Ω source can be achieved by a series inductor and a parallel capacitor.

Figure shows an example of a loaded stub PCB antenna with matching components.

The series inductor and the parallel capacitor transform the antenna impedance to 50 Ω, the input impedance of the filter (FIL1).

For dipole or monopole antennas, the component values for the series inductor and the parallel capacitor (CP) have to be determined by measuring the feed impedance at point A in Figure (with LS = 0-Ω resistor and CP left unpopulated) with a vector network analyzer.

Once this is determined, we can use a Smith Chart to assist in matching the antenna to 50 Ω using LS and CP.

Derivatives of the monopole are the inverted-L and inverted-F antennas as shown in Figure.

In the inverted-L antenna, the monopole does not run perpendicularly to the ground plane over its whole length but is bent parallel to the ground plane after some distance. This helps to save space, but decreases the radiation resistance because the radiator comes closer to the ground plane. An additional matching circuit is needed to match the low-feed impedance to the usual transmission line impedance of 50 Ω.

If we proceed from the feed point of the inverted-L antenna to the end, we notice that the voltage increases (while the current decreases) from a maximum voltage value at the feeding point to almost zero at the end. This means, that the antenna impedance has its minimum if we feed the antenna as shown in Figure 11a) and increases if we move the feeding point towards the end. The inverted-F antenna in Figure 11b) is an inverted-L antenna with a feeding tap that gives larger antenna impedance. If the antenna is tapped at the right location, no additional matching circuit is required.

The structure of inverted-F antennas and in particular the location of the tap is usually determined by electromagnetic simulations.

Transversal Mode Helical Antenna

Another option to reduce the size of a monopole is to coil it up into a helix as shown in Figure.

When the coil circumference and the spacing between adjacent turns are comparable to the wavelength, the antenna radiates a circular polarized beam in the axis of the helix. These antennas are called axial mode helicals.

In small short-range applications, the helix diameter and the spacing between turns are much smaller than a wavelength; the result is a normal mode helical antenna. The radiation pattern of a normal mode helix is similar to that of a monopole; the maximum radiation occurs perpendicular to the helix axis. Due to the shape and the size of the ground plane, radiation patterns of practical antennas can show deviations from this idealized form. The radiation from a normal mode helix is elliptically polarized; usually the component having the electrical field vector parallel to the antenna axis is stronger than the component which is parallel to the ground plane.

The exact calculation of transversal mode helical antennas is not as simple as for dipole and monopole antennas. Usually they are designed empirically: Start with a wire that is half a wavelength long, wind it up into a helix, and measure the antenna impedance using a vector network analyzer. Then cut it back until nearly real input impedance at the frequency of operation is obtained. Real input impedance means that the antenna is in resonance.

Fine-tuning of the frequency of resonance is possible by compressing or stretching the helix. Even if the antenna is in resonance, it will not be matched to 50 Ω yet. The input impedance will be the sum of the radiation and loss resistances, usually smaller than 50 Ω. For the design of the needed additional matching circuit. Chu’s and Wheeler’s limit on the bandwidth for a given dimension also holds for helix antennas.

A small transversal mode helix therefore has tight bandwidth and is sensitive to tolerances of the matching components.

Small Loop Antennas

The loop antenna shown in Figure 13 has a differential feed. Often a ground plane is made part of the loop, giving a single-ended feed as shown in Figure 14.

The small arrows indicate the current flow through the loop. On the ground plane, the current is mainly concentrated on the surface. The electrical behavior of the structure in Figure 14 is therefore similar to that of the loop with differential feed shown in Figure 13.

The following considerations on small loop antennas are based on /4/ and assume that the current is constant over the loop. This means that the circumference must be smaller than one tenth of a wavelength. Although this is rarely the case, the given formulas describe the principal behavior and can be used as a starting point for the loop antenna design.

If the current is constant over the loop, we can consider the loop as a radiating inductor with inductance L. Where L is the inductance of the wire or PCB trace. Together with the capacitor C, this inductance L builds a resonant circuit. Often a resistor Ratt is added to reduce the quality factor of the antenna and to make it less sensitive to tolerances. Of course, this resistor dissipates energy and reduces the antenna’s efficiency.

The following calculations hold for circular loops with the radius a for square loops with the side length a. A rectangular antenna with the sides a1 and a2 will be approximated by an equivalent square with the side length a= sqrt(a1a2). The length (circumference) of the wire building the loop will be called U, where U = 2πa for a circular loop or 4a for a square loop.

For the calculation of the inductance, the wire radius b, where b is 1/2 the diameter of the actual wire used to fabricate the antenna, is needed. In the frequent case where a loop antenna is realized by a trace on a PCB, b = 0.35.d + 0.24.w can be used, where d is the thickness of the copper layer and w is the trace width.

Figure 15 shows the equivalent schematic of a small loop antenna.

The radiation resistance of loop antennas is small, typically smaller than 1 Ω. The loss resistance Rloss describes the ohmic losses in the loop conductor and the losses in the capacitor (expressed by its ESR). Usually, the ESR of the capacitor can not be neglected. Interestingly, the thickness of the copper foil is not needed for the calculation of the loss resistance because due to the skin effect, the current is confined on the conductor surface.

Together with the loop inductance L, which is the inductance of the wire, the capacitor C builds a series resonant circuit. In practice, the L-to-C ratio of this resonant circuit is large giving a high quality factor Q. This would make the antenna sensitive to tolerances. That’s why often an attenuating resistor Ratt is added to reduce the Q. To describe the influence of Ratt on the loop antenna, we make a parallel to series conversion and use the equivalent series resistance Ratt_trans. The resistance value of Ratt_trans is determined by the acceptable tolerance of the capacitor and the geometry of the loop.

The maximum usable quality factor is calculated from the capacitance tolerances ΔC/C:

In most cases, the radiation resistance is much smaller than the loss resistance and the transformed attenuation resistance, giving a poor efficiency. In this case the approximation:

can be used. Rr is determined by the loop area, which is πa2 for circular loops, a2 for square loops, and a1a2 for rectangular loops. Figure 16 shows the efficiency of small circular loop antennas versus their diameter for an assumed tolerance of 5%. The trace width has been assumed as 1 mm, the copper thickness as 50 μm; but both values have only a minor influence on the efficiency, which is mainly determined by the attenuation resistance Ratt. As expected, the efficiency increases with increasing diameter.

If we feed the loop antenna as shown in Figure 14, the series equivalent circuit of Figure 15 describes the antenna impedance. Even including the effect of Ratt, the total series resistance will be small, usually below 10 Ω. If we feed the antenna directly at the capacitor, the parallel equivalent circuit describes the antenna impedance. A small series loss resistor transforms into a large parallel resistor, usually several kΩs.

In both cases matching to 50 Ω will be difficult. That’s why the loop antenna is often tapped, giving an impedance in between the too small and the too large values. Figure 17 shows an example.

A series feed (in the lower right corner) would give a small impedance; a parallel feed (directly at the capacitors) would give a much too large impedance. The tap provides an impedance close to 50 Ω in this example. The loop capacitor has been split into two series capacitors C1 and C2. This makes it possible to realize non-IEC capacitance values. R1 is the damping resistor which de-Q’s the circuit, thus increasing the bandwidth and subsequently reducing the tolerance requirements.

Unfortunately, there are no easy formulas that describe the tapped structure and give the right location for the tap. The line from the antenna termination to the tap is not a transmission line and will disturb the field in the antenna. Therefore, we have to find out the optimal structure by electromagnetic simulations. Often a trial and error procedure is used as an alternative: Using a vector network analyzer we determine the capacitance value that gives the best return loss and the largest resistance value that gives the required bandwidth.

The loop antenna gives a linear polarization with the vector of the electric field oscillating in the plane built by the loop.

As opposed to all the antennas discussed so far, loop antennas are magnetic antennas. This means that they are not detuned by the dielectric constant of the material in their reactive near field. That’s why loop antennas are often used for body-worn or hand-held equipment. Table 2 shows a feature comparison of the discussed antennas.

Rules of Thumb for the Antenna Design

We can summarize the considerations made so far in the following rules of thumb:

• If the available space is sufficient, use a half-wave dipole (for differential feed) or quarter-wave monopole (for single-ended feed) antenna for best efficiency.
• If possible, keep the space around the antenna clear from conducting or dielectric materials, such as electronic components, the casing or the user’s body.
• Sometimes, dielectric materials in the reactive near field are unavoidable. In these cases, measure the antenna impedance under real application conditions and match it to the needed characteristic impedance.
• Due to space limitations, the ground plane of quarter-wave monopoles is often too small. In these cases try to create as much ground plane around the feed point as possible, measure the resulting antenna impedance and match it to the needed characteristic impedance. Good performance can be obtained from a counterpoise made from a quarter wavelength conductor that is connected to ground in the vicinity of the antenna’s feeding point. The counterpoise should run as long as possible perpendicular to the monopole radiator.
• When using premanufactured antennas keep in mind that their performance depends on the attached ground plane. The manufacturer’s specifications are only achieved if the ground plane has the same size and shape as the manufacturer’s evaluation board. In all other cases, you have to measure the impedance of the premanufactured antenna under application conditions and to match it to the needed characteristic impedance.
• Small loop antennas are insensitive to varying dielectric conditions in their reactive near field. They can be a good solution for portable and hand-held devices but have a much lower efficiency than electrical antennas. Only antennas with a circumference smaller than one tenth of a wavelength can be considered as purely magnetic antennas. Larger loops have a higher gain but also a higher sensitivity to the environmental conditions.
• Size matters: Always keep in mind that Chu’s and Wheeler’s limit determines the product of the bandwidth and the efficiency for a given antenna dimension. An extremely small antenna can not be efficient and tolerance-insensitive at the same time.